i) Field of the Invention
The present invention relates to a carrier phase estimation apparatus for estimating a carrier phase of a PSK-modulated received signal in a radio communication field by a movable body such as a mobile satellite communication, and more particularly to a carrier phase estimation apparatus constituted by digital circuits, which is capable of following up a received carrier phase variation caused by fading.
ii) Description of the Related Arts
FIG. 7 illustrates a conventional digital carrier phase estimation apparatus for estimating a carrier phase of a PSK-(phase shift keying-) modulated received signal, as disclosed in "Nonlinear Estimation of PSK-Modulated Carrier Phase with Application to Burst Digital Transmission" by Andrew J. Viterbi and Audrey M. Miterbi, IEEE Transactions of Information Theory, Vol. IT-29, No. 4, pp. 543-551, July, 1983. As shown in FIG. 7, a base band converter 1 takes a received IF signal 3 as input and outputs an in-phase component In and an quadrature component Qn of a digital base band signal. A phase estimator 2 receives the in-phase component In and the quadrature component Qn output from the base band converter 1 and estimates a received carrier phase .theta. every symbol and outputs a carrier phase .theta. estimated every symbol (herein-after referred to as an estimated carrier phase .THETA.n; n=0, 1, 3, . . . ). The base band converter 1 includes a pair of multipliers 4 and 5 for receiving the received IF signal 3 and the output signals of the multipliers 4 and 5 are supplied to a pair of LPFs (low pass filters) 8 and 9, respectively. The output signals of the LPFs 8 and 9 are fed to a pair of A/D (analog-digital) converters 10 and 11, respectively, and the A/D converters 10 and 11 convert the output signals into digital signals to obtain the in-phase component In and the quadrature component Qn of the digital base band signal. The base band converter 1 further includes a carrier oscillator 8 and a 90.degree. phase shifter 7. Because of the carrier oscillator 6 and the 90.degree. phase shifter 7, carrier signals having 90.degree. different phases are supplied to the multipliers 4 and 5.
The phase estimator 2 includes a non-linear circuit 12 which receives the in-phase component In and the quadrature component Qn of the digital base band signal and executes a non-linear operation. The in-phase and quadrature components In' and Qn' output by the non-linear circuit 12 are input to a pair of filters 13 and 14 and the output signals of the filters 13 and 14 are sent to a coordinate transformer 18 which transforms the output signals into a carrier phase .THETA.n. The filters 13 and 14 have the same construction and each filter 13 or 14 is constituted by N (N is a positive integer) stages of shift registers 15, an adder 16 and a divider 17.
Next, an operation of the conventional carrier phase estimation apparatus shown in FIG. 7 will be described in detail.
First the received IF signal 3 is converted into the in-phase component In and the quadrature component Qn of the digital base band signal by the base band converter 1. In the base band converter 1, the received IF signal 3 is branched off in two directions, that is, one branch is input to the multiplier 4 and the other branch is input to the multiplier 5. The carrier oscillator 6 outputs a carrier having the same frequency as that of the received IF signal 3 and the output signal of the carrier oscillator 6 is supplied to the multiplier 4. As a result, in the multiplier 4, a product of the received IF signal 3 and the output signal of the carrier oscillator 6 is obtained. On the other hand, the output signal of the carrier oscillator 6 is input to the multiplier 5 via the 90.degree. phase shifter 7. As a result, in the multiplier 5, a product of the received IF signal 3 and the output signal of the 90.degree. phase shifter 7 is obtained.
The output signal of the multiplier 4 is input to the LPF 8 and the LPF 8 removes unnecessary harmonic components from the output signal of the multiplier 4 and outputs the in-phase component of the base band signal. The output signal of the multiplier 5 is input to the LPF 9 and the LPF 9 removes unnecessary harmonic components from the output signal of the multiplier 5 and outputs the quadrature component of the base band signal. The in-phase component of the base band signal output from the LPF 8 is input to the A/D converter 10 in which the in-phase component is sampled by a clock (not shown) of a symbol period and is converted into the in-phase component In of the digital base band signal, and the A/D converter 10 outputs the in-phase component In of the digital base band signal. The quadrature component of the base band signal output from the LPF 9 is input to the A/D converter 11 in which the quadrature component is sampled by a clock (not shown) of a symbol period and is converted into the quadrature component Qn of the digital base band signal, and the A/D converter 11 outputs the quadrature component Qn of the digital base band signal.
In this case, for simplifying and better understanding of the explanation, it is assumed that the received IF signal 8 is a BPSK-(binary PSK-) modulated signal with no noise and a phase difference between the received IF signal 3 and the output signal of the carrier oscillator 6 is, for example, 22.5.degree.. For example, when a data series is "10110", the in-phase component In and the quadrature component Qn of the digital base band signal are represented by formulas (1) represented hereinbelow. And these components are shown by (a) to (e) in FIG. 8. EQU In=A.multidot.d.sub.n .multidot.cos.theta.n EQU Qn=A.multidot.d.sub.n .multidot.sin.theta.n (1)
In the formulas (1), A represents an amplitude and dn represents a value of either +1 or -1 corresponding to "1" and "0". In this case, the value of 22.5.degree. is a value of the carrier phase to be estimated by the carrier phase estimation apparatus.
Next, the in-phase component In and the quadrature component Qn of the digital base band signal are input to the phase estimator 2. In the phase estimator 2, first, the components In and Qn are input to the non-linear circuit 12 which executes a non-linear operation represented by formula (2) shown below against the in-phase component In and the quadrature component Qn. EQU In'=.rho..multidot.cosm.theta.n EQU Qn'=.rho..multidot.sinm.theta.n (2)
In the formulas (2), m represents a value corresponding to an m-phase PSK signal. For instance, in the case of the BPSK signal, because of binary phase, m=2 and in the case of a QPSK signal, m=4. Also, .rho. possesses a function for changing the estimation characteristics of the carrier phase by its value. In the aforementioned document, it is disclosed that good results can be obtained when .rho.=1 or .rho.=In.sup.2 +Qn.sup.2. In this case, for simplifying the explanation, assuming that .rho.=1, when the received IF signal 3 is the BPSK-modulated signal, the output signals In' and Qn' of the non-linear circuit 12 are shown by formulas (3) as follows. EQU In'=cos2.theta.n EQU Qn'=sin2.theta.n (3)
FIG. 9 illustrates the signals In' and Qn' which are obtained by modifying the in-phase component In and the quadrature component Qn of the digital base band signal shown in FIG. 8 by the non-linear circuit 12. In this case, 2.theta..sub.1 =2.theta..sub.2 =2.theta..sub.3 =2.theta..sub.4 =2.theta..sub.5 =45.degree. and the data modulating component is removed. That is, the non-linear circuit 12 removes the data modulating component of the m-phase PSK signal.
FIG. 10 shows the above-described converted values In' and Qn' as a list. As shown in FIG. 10, in this case, the in-phase components of the digital base band signal become In'=cos2.theta.n=cos45.degree.=0.707 and the quadrature components of the same become Qn'=sin2.theta.n=sin45.degree.=0.707.
The in-phase components In' and the quadrature components Qn' of the digital base band signals, modified and output by the non-linear circuit 12, are supplied to the respective filters 13 and 14 in which the noise included in the components In' and Qn' is reduced.
In the filter 13, the in-phase components In' of the digital base band signal is input to the shift registers 15. The N number of components In' input in the shift registers 15 are added to each other in the adder 16. At this time, the value of the symbol number n of the filter output signal Xn is equal to the symbol number n of the component In' input in the central stage of the shift registers 15. For example, as shown in FIG. 10, in the state that 5 components In' of n=1 to 5 are input in the shift registers 15, the component I.sub.3 ' is input in the central stage of the shift registers 15 and thus the filter output signal Xn is X.sub.3. That is, the filter 18 performs the operation for taking an average by using the (N-1)/2 numbers off the components In' aligned in the front and rear sides of the central component In' input in the central stage of the shift registers 15 to reduce the noise.
The filter 14 has the same construction as the filter 13 and operates the input quadrature components Qn' of the digital base band signals in the same manner as the filter 13 to output a filter output signal Yn.
In the example shown in FIG. 10, ##EQU1##
Next, the filter output signals Xn and Yn are input to the coordinate transformer 18. The coordinate transformer 18 executes an operation shown in formula (4) and outputs the estimated carrier phase .THETA.n. EQU .THETA.n=(1/m).multidot.tan.sup.-1 (Yn/Xn) (4)
In formula (4), m represents the value corresponding to the m-phase PSK signal, and in the case of the BPSK signal, m=2. Also, in the case of the QPSK signal, m=4. In the example shown in FIG. 10, it is calculated as follows. EQU .THETA.n=(1/2).multidot.tan.sup.-1 (Y.sub.3 /X.sub.3)=22.5.degree.(5)
In this case, the true estimation value of the carrier phase .THETA.=22.5.degree. is estimated. The estimation operation of the above-described carrier phase is carried out against each symbol (n=. . . , -2, -1, 0, 1, 2, 3, . . . ).
In the conventional embodiment described above, since the case that no noise is attached to the received IF signal 3 is described, the carrier phase can be estimated without any error. However, in case that noise Is contained in the received IF signals 3, in order to remove the influence of the noise, a required larger number of filter stages are designed.
In FIG. 11, there is shown a conventional data demodulation apparatus using the carrier phase estimated in the above-described carrier phase estimation apparatus.
As shown in FIG. 11, a sine wave generator 19 inputs the estimated carrier phase .THETA.n output from the phase estimator 2 and outputs cosine and sine signals of cos.THETA.n and sin.THETA.n. A pair of multipliers 20 and 21 multiply the output signals cos.THETA.n and sin.THETA.n of the sine wave generator 19 by the in-phase component In and the quadrature component Qn of the digital base band signal, respectively, to output signals t.sub.n and u.sub.n, respectively. The output signals t.sub.n and u.sub.n of the multipliers 20 and 21 are added to each other in an adder 22. The added value of the adder 22 is supplied to a discriminator 23 and the discriminator 23 discriminates the input data and outputs demodulated data.
The in-phase component In and the quadrature component Qn of the digital base band signal are represented by formulas (1) and hence the signals t.sub.n and u.sub.n output from the multipliers 20 and 21 are represented in formulas (6) as follows. EQU t.sub.n =A.multidot.d.sub.n .multidot.cos.theta.n.multidot.cos.THETA.n EQU u.sub.n =A.multidot.d.sub.n .multidot.sin.theta.n.multidot.sin.THETA.n (6)
When the carrier phase estimation apparatus correctly estimates the carrier phase, .THETA.n=.theta.n and the formulas (6) can be rewritten to formulas (7) as follows. EQU t.sub.n =A.multidot.d.sub.n .multidot.cos.sup.2 .theta.n EQU u.sub.n =A.multidot.d.sub.n .multidot.sin.sup.2 .theta.n (7)
As a result, the output signal of the adder 22 is t.sub.n +u.sub.n =A.multidot.d.sub.n. Hence, the discriminator 23 discriminates the positive and the negative of the output signal A.multidot.d.sub.n of the adder 22 to demodulate the data.
As described above, the conventional carrier phase estimation apparatus can be realized by using the digital signal processing technique and thus is suitable for miniaturization and non-adjustment. However, when the carrier phase estimation apparatus is required to be mounted on a mobile or movable body, for example, in a mobile communication system or a mobile satellite communication system by which a movable body performs communication by using satellites, the received carrier phase is affected by fading and thus is usually fluctuated.
Accordingly, when the conventional carrier phase estimation apparatus is applied to such communication systems as it is, the carrier estimation characteristics are degraded by the fading and bit error rate characteristics of the demodulated data become deteriorated. This problem will be described in connected with embodiments.
FIG. 12 illustrates the in-phase components In and the quadrature components Qn of the digital base band signal in a similar manner to the example of the BPSK signal shown in FIG. 8. However, in the instance shown in FIG. 12, even when no noise is attached to the received IF signal 3, the in-phase component In and the quadrature component Qn of the digital base band signal are fluctuated by the influence of the fading.
In this shown example, the carrier phase of the first symbol (n=1) is 22.5.degree., the carrier phase of the second symbol (n=2) increases 12.5.degree. compared to the carrier phase of the first symbol, the carrier phase of the third symbol (n=3) increases 15.degree. compared to the carrier phase of the second symbol, the carrier phase of the fourth symbol (n=4) decreases 15.degree. compared to the carrier phase of the third symbol, and the carrier phase of the fifth symbol (n=5) decreases 12.5.degree. compared to the carrier phase of the fourth symbol. That is, the carrier phases fluctuate as follows.
.theta..sub.1 =22.5.degree. PA1 .theta..sub.2 =215.degree. PA1 .theta..sub.3 =50.degree. PA1 .theta..sub.4 =35.degree. PA1 .theta..sub.5 =202.5.degree. PA1 2.theta..sub.1 =45.degree. PA1 2.theta..sub.2 =70.degree. PA1 2.theta..sub.3 =100.degree. PA1 2.theta..sub.4 =70.degree. PA1 2.theta..sub.5 =45.degree. PA1 I.sub.1 '=cos2.theta..sub.1 =cos45.degree.=0.707 PA1 I.sub.2 '=cos2.theta..sub.2 =cos70.degree.=0.342 PA1 I.sub.3 '=cos2.theta..sub.3 =cos100.degree.=-0.174 PA1 I.sub.4 '=cos2.theta..sub.4 =cos70.degree.=0.342 PA1 I.sub.5 '=cos2.theta..sub.5 =cos45.degree.=0.707 PA1 Q.sub.1 '=sin2.theta..sub.1 =sin45.degree.=0.707 PA1 Q.sub.2 '=sin2.theta..sub.2 =sin70.degree.=0.940 PA1 Q.sub.3 '=sin2.theta..sub.3 =sin100.degree.=0.985 PA1 Q.sub.4 '=sin2.theta..sub.4 =sin70.degree.=0.940 PA1 Q.sub.5 '=sin2.theta..sub.5 =sin45.degree.=0.707
FIG. 13 shows the output signals In' and Qn' of the non-linear circuit 12 corresponding to the in-phase component In and the quadrature component Qn of the digital base band signal shown in FIG. 12. That is, the following values are obtained.
FIG. 14 illustrates the output signals In' and Qn' shown in FIG. 13 by numerical values. That is, relating to the in-phase components In', the following values are obtained.
Also, regarding the quadrature component Qn', the following values are obtained.
Hence, the filter output signals Xn and Yn of the filters 13 and 14 are obtained as follows. ##EQU2## Thus, the estimated carrier phase .THETA..sub.3 is calculated as follows. EQU .THETA..sub.3 =(1/2).multidot.tan.sup.-1 (0.856/0.385)=32.9.degree.
Since the true value of the carrier phase to be estimated is .theta..sub.3 =50.degree., the carrier phase .THETA..sub.3 =32.9.degree. actually estimated by the above-described conventional carrier phase estimation apparatus has an estimation error of 17.1.degree. from the true value. This estimation error is caused due to the fact that, though the carrier phases fluctuate due to the influence of fading, the average value is simply taken without considering any fluctuation of the carrier phases in the conventional carrier phase estimation apparatus. That is, in the conventional carrier phase estimation apparatus, only the noise is considered and a simple average value is taken for removing the influence of the noise. Hence, when the conventional carrier phase estimation apparatus is used for mobile communication systems in which fading is generated, the previously remarkable bit error rate characteristics are degraded and it is difficult to apply the conventional carrier phase estimation apparatus to such communication systems as it is.
As described above, the conventional carrier phase estimation apparatus does not include any good follow-up characteristics against the carrier phase fluctuation caused by fading and thus the bit error rate characteristics in a fading channel are deteriorated.